Voltage Regulated Single Switch DC-DC Converter with High Voltage gain

Voltage Regulated Single Switch DC-DC Converter with High Voltage gain

Susmi. K. M, Meera. G, Remya. M. S

PG Students(Power Electronics),EEE Dept Vidya Academy Of Science And Technology Kerala, India

AbstractThis paper develops a voltage regulated single switch DC DC converter with high voltage gain. The proposed converter is a non isolated structure using coupled inductor technique. For applications like uninterrupted power supply, electric traction etc a high voltage gain is required. In this paper, a high voltage gain is obtained by the coupled inductor and the diode-capacitor combination at the secondary side of the coupled inductor. The output voltage is regulated to obtain a constant value for small variations in the input. The clamp circuit connected to the primary winding of the coupled inductor clamps the voltage across the switch to a lower value. This enables the use of low voltage rating power devices. Efficiency increases as the leakage inductance energy is recycled.

Keywordscoupled inductor, high voltage gain, voltage regulated

1. INTRODUCTION

   We cannot imagine a world devoid of power. The need for power is thriving each day. The excessive use of fossil fuels has resulted in their depletion and various environmental problems. The energy from renewable resources needs to be depended more in the future.

   For applications like uninterruptable power supply, photovoltaic generation system, electric traction etc a conventional boost converter is used for stepping up the input voltage. The drawback of this converter is high voltage stress across the switches and diodes. The voltage stress is equal to the output voltage. This increases the switching losses. So higher voltage rating switches have to be used. The efficiency of the conventional boost converter decreases with increase in duty ratio. So various voltage boosting topologies were introduced.

   Switched capacitor based converters [2] – [3] consist of switches and capacitors only. Similarly switched inductor type converters [4] – [5] consist of switches and inductors. The drawback with these converters is the requirement of large number of switches and capacitors/ inductors for obtaining high voltage gain. This makes the circuit more complex and costly.

   Non isolated converters like flyback converter, coupled inductor based converters were also used. The turns ratio of the transformer is adjusted in flyback converter to obtain high voltage gain. But the leakage inductance of transformer cause voltage spikes and increases the losses. Coupled inductor converters provide high voltage gain by compromising on the duty ratio.

   In this paper, a single switch DC DC converter with high voltage gain is presented. The voltage gain of the converter is increased by the coupled inductor. The voltage gain is further increased by the diode- capacitor combination connected to the secondary winding of the coupled inductor. The output voltage is regulated to provide a constant voltage at the output for small variation in the input voltage. The voltage across the switch is clamped to a low voltage value by the clamp circuit connected to the primary winding of the coupled inductor. Hence, low voltage rating switches can be used. The leakage inductance energy of the coupled inductor is recycled and hence an increase in the efficiency can be observed.

   The outline of the paper is as follows:

   Section II deals with the overall system configuration. This includes the circuit structure, equivalent circuit and assumptions for the analysis of the circuit. Modes of operation of the circuit is discussed in section III. The converter operates in five modes. Section IV is the steady state analysis. The voltage gain of the circuit, the voltage stress across the switches and diodes are derived in this section. Section V provides the simulation results. The simulation diagram, simulation parameters and simulation waveforms are provided. Conclusions from the results are provided in the section VI.

2. OVERALL SYSTEM CONFIGURATION

   This section covers the circuit structure, the simplified equivalent circuit and assumptions taken into consideration for circuit analysis. The circuit structure is as in Fig.1. It consists of a single switch Q, coupled inductor T1, input inductor L1, A clamp circuit with capacitor C3 and diode D3, voltage doubler cell with diodes D3 and Dr and capacitor C3. Dr is regeneration diode. The circuit has an output capacitor C0, diodes D1, D2,

   D0. The couple inductor is modeled as an ideal transformer with N (n2/n1)as turns ratio. A parallel magnetizing inductance Lm and leakage inductance LK1 and LK2 are shown

   in the equivalent circuit Fig.2.

   The assumptions considered for circuit analysis are as follows:

   • All components are ideal except the leakage inductance of the coupled inductor T1.

   • The input inductance L1 is assumed to be sufficiently large so that the current through L1 is continuous.

   • The capacitors are assumed to be sufficiently large so that the voltage across the capacitors is constant

   • The converter is working under continuous conduction mode. Both the inductors L1 and Lm operate under continuous conduction mode.

   • The duty ratio should be less than 0.7.

   capacitors C1, C2, C3, Co are represented as VC1, VC2, VC3, VC0 respectively. The voltage across the output resistor R0 is V0 and the current through R0 is i0. The current through capacitor

   C0 is iC0 and that across C3 and C2 is iC3 and iC2 respectively. The voltage across the magnetizing inductance Lm is VLm. The current through the leakage inductance LK1 and LK2 are iLK1 and iLK2 respectively.

3. MODES OF OPERATION

   The circuit operates under continuous conduction mode. There are five modes of operation for the circuit. The waveforms of the converter are shown in fig.8

   A. Mode 1 ( t0 – t1 )

   Fig. 1 Circuit structure for the proposed converter

   Fig. 3 shows the equivalent circuit for Mode 1 operation of the converter. In this mode, at t = t0, the switch Q is turned ON. The diodes D2, Dr are forward biased and D1, D3, D0 are reverse biased. The voltage across the diodes D1, D3 and D0 are VC1, VC1 + VC2 and V0 VC1 VC2 respectively. The current flow path is shown in Fig. 3 with dotted arrows. The

   Fig. 2 Equivalent circuit of the proposed converter

   input voltage is transferred to the inductor L1 through D2 and

   In Fig. 2, input voltage is represented as Vin, the current through inductor L1 as IL1and voltage as VL1. The voltage across the diodes D1, D2, D3, Dr, Do are represented as VD1, VD2, VD3, VDr, VDo respectively. The voltage across the

   Q. The inductor charges and hence the inductor current iL1 increases linearly. The capacitor C1 discharges through the magnetizing inductance Lm, leakage inductance LK1 and he switch Q. So the primary voltage of the coupled inductor is

   VC1. The capacitors C1 and C2 discharges through diode Dr and charges C3. Output power is supplied by the capacitor C0.

   Fig. 3 Equivalent circuit for Mode 1 operation of the proposed converter

   1. Mode 2 ( t1 t2 )

      Fig 4 shows the equivalent circuit for mode2 operation . At t = t1, the switch Q is turned OFF. The diode D1, D3 and Dr are forward biased. Diodes D2 and D0 are reverse biased. The voltage across diode D2 is VC2 and that across diode D0 is V0 VC1 – VC2.The current through Q in stage 1 is forced to flow through D3 in this stage. The inductor L1 discharges through the diode D1 and charges the capacitor C1. The inductor current iL1 decrases linearly. The energy stored in LK1 discharges through the diode D3 and charges the capacitor C2. Here, the energy stored in inductor LK1 is recycled to capacitor C2. The energy stored in inductor LK2 discharges through the diode D3 and Dr and charges the capacitor C3. The output power is supplied by the capacitor C0. The voltage stress across the switch Q is VC1 + VC2. At t = t2, the inductor current iLK2 reaches zero.

      Fig.4 Equivalent circuit for Mode 2 operation of the proposed converter

   2. Mode 3 ( t2 t3 )

      Fig.5 shows the equivalent circuit for mode 3 operation. At t = t2, the inductor current iLK2 reaches zero and the switch Q remains in OFF state. Diodes D2 and Dr are reverse biased and D1, D3 and D0 are forward biased. The voltage across diode D2 is VC2 and that across diode Dr is VC3 + NVC2. The inductor L1 continues discharging through the diode D1 and charges the capacitor C1. The inductor current iL1 keeps on decreasing linearly. The energy stored in LK1 discharges through the diode D3 and charges the capacitor C2. The energy stored in inductor Lm also discharges through n2 and C3 and provides the output voltage. Here, the leakage inductor energy is recycled. The voltage stress across the switch Q is VC1 + VC2. At t = t3, iLK1 = iLK2 and the current through the capacitor C2 (iC2) is zero.

      Fig. 5 Equivalent circuit for Mode 3 operation of the proposed converter

   3. Mode 4 ( t3 t4 )

      Fig. 6 shows the equivalent circuit for mode 4 operation. The switch Q continue to remain in OFF condition. The diodes D2, D3, Dr are reverse biased. Diodes D1 and D0 are forward biased. The inductor current continue decreasing as the energy stored in inductor L1 discharges through the diode D1 to charge the capacitor C1. The input voltage Vin, inductor L1, Lm, LK1, winding n2, LK2 and C3 are connected in series and this providesthe output power at the load R0. At t = t4, switch Q is turned ON.

   4. Mode 5 ( t4 t5 )

   Fig. 6 shows the equivalent circuit for 4 operation. At t = t4, switch Q is turned ON. Diodes D1, D3, Dr are reverse biased. The voltage across the reverse biased diodes are VC1, VC1 + VC2, V0 VC1 VC2 respectively. The diode D2 and D0 are forward biased. The inductor L1 charges through the diode D1. So the inductor current increases linearly. The capacitor C1 discharges through Lm and LK1. So the current

   through the inductors increases linearly. The magnetizing inductor Lm transfers its energy to the load through the secondary winding. At t = t5, the current through LK2 decreases to zero and iLm = iLK1.

   Fig.6 Equivalent circuit for Mode 4 operation of the proposed converter

   Fig. 7 Equivalent circuit for Mode 5 operation of the proposed converter

4. STEADY STATE ANALYSIS

   A lossless power condition is assumed for the steady state analysis. The leakage inductances of the coupled inductor are neglected. Mode 1 and Mode 3 are only considered for the analysis as the time duration of other modes are very small. From Mode 1, the following equations are obtained.

   Fig. 8 Theoretical waveforms of the proposed converter

   VL1 = Vin (1)

   VLm = VC1 (2)

   VC3 = NVC1 + VC1 + VC2 (3)

   From Mode 3, the following equations are obtained.

   VL1= Vin- VC1 (4)

   VLm= – Vc2 (5)

   V0 = VC1 + (N + 1)VC2 + VC3 (6)

   Using the volt second balance principle to inductor L1 and Lm

   , the following equations are obtained.

   VinDTs+ (Vin- VC1) 1-DTs = 0 (7)

   VC1DTs+ – VC2 1-DTs =0 (8)

   The voltage across the capacitors C1, C2 and C3 can be derived from (1) (8)

   The voltage stress across the diode in the conventional boost converter is V0. The above equations show that the voltage stress across diodes is reduced in the proposed converter.

   VC1

   = Vin (1-D)

   = 1-D V0

   (2+N)

   (9)

   The turns ratio of the coupled inductor T1 can be

   calculated by substituting (9), (10) and (11) in (6)

   VC2

   = DVin

   (1-D)2

   = 0

   (2 + )

   (10)

   as:

   = 0 (1 ) 2 2 (19)

   V = (N+1-DN)Vin = (N+1-DN)V0

   (11)

   C3 (1-D)2

   (2+N)

5. SIMULATION RESULTS

   Substituting (9), (10), (11) in (6), voltage gain is obtaines as :

   M= V0 = (2+N) Vin (1-D)2

   The maximum voltage stress across the switch Q is derives as:

   (12)

   The proposed converter was simulated using Matlab Simulink 2009b. The converter was designed and simulated for an output power of 50Watt. The input voltage is 10V and the converter generates an output of 120Volt. In order to regulate the output voltage a PI controller is used. The output from the converter is measured and compared with a constant voltage of 120Volt. The error is given to a PI controller to reduce the steady state error. The output from the saturation block is compared with a saw tooth waveform to generate the switching pulses.

   Vmax

   Q= V0 (2+N)

   (13)

   A. Simulation Parameters

   The values for various components are as given in Table.1.

   Comparing the conventional boost converter with the proposed converter, the voltage stress across the switch is reduced . The voltage stress across the switch in conventional boost converter is V0. In the proposed converter the voltage stress across the switch decreases as turns ratio increases.

   The voltage stress across the diodes are obtained as:

   The output voltage remains at 120 Volt for small variations in the input voltage. Hence voltage regulation is attained. Fig. 9 shows the simulation diagram.

   Vstress

   D1= VC1

   = Vin (1-D)

   = (1-D)V0

   (2+N)

   (14)

   Vstress

   D2=VC2

   = DVin

   (1-D)2

   = DV0 (2+N)

   (15)

   Vstress D3

   = VC1

   + VC2

   = Vin

   (1 D)2

   = V0

   (2 + N)

   VstressD0 = V0- VC1- VC2 =

   (16)

   (1+N)Vin = (1-D)2

   (1+N)V0 (17) (2+N)

   Fig.9 Simulation Diagram

   The output voltage waveform is as in Fig. 10. For an input voltage of 10 volt an output of 120 volt is obtained. Fig. 11 shows the switching pulse applied to the switch Q. The voltage and current through the switch is as shown in Fig.12.

   = V0- VC1

   – VC2

   = (1 + )

   (1 )2

   = (1 + )0

   (2 + )

   (18)

   Fig.13 shows the inductor current iL1. The current through primary and secondary leakage inductor is shown in Fig 14 and Fig. 15 respectively. The capacitor currents iC1, iC2, iC0 are shown in Fig. 16, Fig. 17, Fig.18 respectively. The voltage stress across the diodes is shown in Fig.19.

   TABLE 1. Simulation parameters

   Simulation parameters	Values
   Output power	50 W
   Vin	10V
   V0	120V
   fs	40kHz
   N = n2/n1	2
   Lm	370µH
   L1	60µH
   C1	470µF
   C2	47µF
   C3	47µF
   C0	470µF

   Fig. 10 Output Voltage Waveform

   Fig. 11 Switching pulses to the switch Q

   Fig. 12 Voltage and current waveforms across switch Q

   Fig. 13 Current through the inductor L1

   Fig. 14 Current through primary leakage inductor LK1

   Fig. 15 Current through secondary leakage inductor LK2

   Fig. 16 Current through the capacitor C1

   Fig. 17 Current through the capacitor C2

   Fig.18 Current through the capacitor C0

   Fig.19 Voltage stress across the diode D1, D2, D3, Dr, D0

6. CONCLUSION

A single switch DC DC converter with high voltage gain is presented in this paper. The voltage gain of the converter is increased by the coupled inductor. The voltage gain is further ncreased by the diode- capacitor combination connected to the secondary winding of the coupled inductor. The output voltage is regulated to provide a constant voltage at the output for small variation in the input voltage. The voltage across the switch is clamped to a low voltage value by the clamp circuit connected to the primary winding of the coupled inductor. Hence, low voltage rating switches can be used. The leakage inductance energy of the coupled inductor is recycled and hence an increase in the efficiency can be observed. The converter was simulated using Matlab Simulink 2009b and the output waveforms obtained were as per the theoretical values. With 50 Watt power, an input of 10 Volt produces an output of 120 Volt with a switching frequency of 40 kilo Hertz. The voltage stress across the switch is 30 Volt. This shows that the voltage stress across the switch is reduced comparing with conventional boost converter. The voltage stress across the diodes are also reduced in the proposed converter.

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